Millimeter-Wave Integrated-Circuit Antenna Arrays Citation Tong, Peter Ping Tak (1985) Millimeter-Wave Integrated-Circuit Antenna Arrays. Dissertation (Ph.D.), California Institute of Technology. doi:10.7907/ddxq-rj80. https://resolver.caltech.edu/CaltechETD:etd-08172005-102232 Abstract This thesis presents three different types of millimeter-wave integrated-circuit antenna array. They are a linearly polarized antenna array that can form polarization and intensity line images, a circularly polarized spiral antenna and a two-dimensional tracking antenna array. They are all integrated-circuit antennas with detectors on quartz substrates using quasi-optical systems to collect and focus the waves. Item Type: Thesis (Dissertation (Ph.D.)) Subject Keywords: Electrical Engineering Degree Grantor: California Institute of Technology Division: Engineering and Applied Science Major Option: Electrical Engineering Thesis Availability: Public (worldwide access) Research Advisor(s): Rutledge, David B. Thesis Committee: Rutledge, David B. (chair) Bellan, Paul Murray Bridges, William B. Knowles, James K. Papas, Charles Herach Psaltis, Demetri Defense Date: 12 October 1984 Funders: Funding Agency Grant Number Caltech UNSPECIFIED Record Number: CaltechETD:etd-08172005-102232 Persistent URL: https://resolver.caltech.edu/CaltechETD:etd-08172005-102232 DOI: 10.7907/ddxq-rj80 Default Usage Policy: No commercial reproduction, distribution, display or performance rights in this work are provided. ID Code: 3155 Collection: CaltechTHESIS Deposited By: Imported from ETD-db Deposited On: 17 Aug 2005 Last Modified: 16 Apr 2021 22:12 Thesis Files Preview PDF (Tong_pp_1985.pdf) - Final Version See Usage Policy. 5MB Repository Staff Only: item control page

MILLIMETER-WAVE INTEGRATED-—CIRCUIT ANTENNA ARRAYS Thesis by Peter P. Tong In Partial Fulfillment of the Requirements for the Degree of Doctor of Philosophy California Institute of Technology Pasadena, California 198 (Submitted October 12, 1984) ii ACKNOWLEDGMENTS This work could never have been done without the help of many. First I would like to thank God for giving me strength in these four years at Caltech when I needed it. Next I would like to thank my advisor, Dave Rutledge. His constant encouragement ,enthusiasm and excellent guidance in the work stimulated me. Also I would like to thank Dean Neikirk for setting up most of the instruments I used and teaching me how to build integrated-circuits. Moreover, I am indebted to Dr. N. C. Luhmann, Jr. and his group at UCLA especially. Peter Young and Tony Peebles for letting me use their instruments and being very helpful. . For making life more interesting, I would like to thank the guys in the Tai-Yung House (Cheung Kwok Wai, Fai Mok, Ho Chi Fai, Alex Ho and Tang Tin Wu ) and the Millimeter-Wave Integrated-Circuit group here at Caltech (Joyce, Chung-en Zah, Wayne Lam, Dayalan Kasilinganm, Rick Compton and Wyman Williams ). Also my thanks to Mr. and Mrs. B. Lee and their family for helping me to settle down when I first arrived in Los Angeles. Financially, I would like to thank Caltech for supporting me. Finally I express my deepest gratitude to my parents, for their unrelentless support. iii MILLIMETER-WAVE INTEGRATED-CIRCUIT ANTENNA ARRAYS ABSTRACT This thesis presents three different types of millimeter-wave integrated-circuit antenna array. They are a linearly polarized antenna array that can form polarization and intensity line images, a circularly polarized spiral antenna and a two-dimensional tracking antenna array. They are all integrated-circuit antennas with detectors on quartz substrates using quasi- optical systems to collect and focus the waves. iv TABLE OF CONTENTS Acknowledgements Abstract 1. Introduction a. b. Millimeter-wave properties and applications Existing technologies c. Overview of the thesis 2. Linearly-polarized antenna array for polarization imaging a. Applications of polarimeters b. ge h. i. Some existing polarimeters from optics to microwaves The linearly polarized antenna array Antenna spacing for diffraction-limited images The design and properties of the antenna array A photoresist shadowing technique to make the array Optical setup Measuring the polarization angle Imaging a step change in polarization Appendix: common definitions in antenna polarization page ii iii wb »b FS 14 16 20 23 23 30 Vv Table of Contents 3. Circularly polarized spiral antenna array ae b. c. d. e. £. Circularly polarized spiral antenna: motivation The equiangular spiral antenna in free space The design and impedance of the spiral on dielectric Integrated-circuit fabrication of the spiral The axial ratio of the spiral on dielectric Antenna pattern measurements at millimeter wavelengths The antenna patterns of the spiral on dielectric The efficiency of the spiral on dielectric 4. Two-dimensional tracking antenna array The idea of the array The design, impedance and antenna patterns of the array Integrated-circuit fabrication of the array The tracking of a target 5. Future work in millimeter waves Appendices: 1. The response of any polarized antenna to any polarized wave ae b. Cc. The linearly, elliptically and circularly polarized antennas The generation of elliptically polarized waves The theoretical and experimental responses 37 37 39 39 42 44 47 52 56 59 59 64 67 68 76 78 78 81 86 2. vi Table of Contents of the antennas to the waves The antenna patterns of a long wire ona dielectric a. The theoretical and experimental H-plane patterns b. The experimental and parameter-fitted E~plane pattern The computer program for antenna pattern measurement 96 96 100 103 1/ INTRODUCTION Millimeter-wave properties and applications Millimeter and submillimeter waves have unique properties that make them superior in many applications. They penetrate smoke and dust with less attenuation than the optical and infrared wavelengths. At the same time millimeter and submillimeter wave systems achieve higher resolution with smaller and lighter weight components than their microwave counterparts. One can foresee applications of these wavelengths for short-range communications [1,2] and satellite communications [3,4,5]. Another use of these wavelengths is in fusion experiments in measuring electron density and magnetic fields. Researchers in the fusion experiments are trying to measure the magnetic field in the plasmas by Faraday rotation. These are the natural wavelengths for these measurements because there are problems with refraction at microwave wavelengths and vibration at infrared wavelengths [6,7]. This technology has also been applied productively in the radioastronomy field. It has opened up a new region of the spectrum for interesting observations [8]. These wavelengths are scattered less than shorter wavelengths, and it is also possible to observe 2 Introduction relatively cool objects such as developing stars in large gas clouds. _ Existing technologies Metal waveguides are used at these wavelengths. They become lossy at higher frequencies (about 6 dB/m in rectangular waveguide at 100 GHz). The waveguide dimensions become small and machining becomes difficult and expensive. Also it is hard to mount matching and coupling structures in the small waveguides. In order to get around these problems, we used quasi-optical systems to collect and focus the waves. Integrated- circuit technology is employed to make the antennas and detectors in receivers. This approach gives a flexible design and is relatively inexpensive. Several millimeter-wave imaging systems have been developed based on a single detector and scanned optics [9,10,11,12]. In many experiments, though, the image may be changing too quickly or there may not be enough integration time for scanned optics. Several workers have proposed imaging arrays [13,14,15], and Neikirk and Rutledge have demonstrated arrays that image intensity at 1.2 mm and 119 pm [16,17]. Overview of the thesis The polarization of an electric field at millimeter wavelengths is important in many studies and applications including distinguishing man-made and natural objects in radar[18], studying plasmas [6], and suppressing rain echoes [19]. Chapter two of this thesis 3 Introduction will present an intensity and polarization imaging System at a wavelength of 0.8 mm using an array of linearly polarized modified bow-tie antennas. Chapter three presents a millimeter-wave circularly polarized _integrated-circuit antenna. In order to form a two- dimensional image with a one-dimensional array. one - Still needs to scan the array in one direction. However, this scanning can be eliminated as well if a two-dimensional array is available. Chapter four of this thesis will show a two-dimensional tracking antenna array on a dielectric. This array is presently undergoing further development as a GaAs integrated- circuit at the Hughes Aircraft Corporation. Three separate topics are covered in the Appendices. In Appendix 1, the response of any polarized antenna to any polarized wave is studied. The experimental results fit well with theory. In Appendix 2, the antenna patterns measured at millimeter wavelengths of a long wire antenna on a dielectric are presented together with theory. The final Appendix contains the source code for the program that controls the antenna-pattern measurement. Part of the work in this thesis has been previously published : 1. P. P. Tong, D. P. Neikirk, D. Psaltis, D. B. Rutledge, K. Wagner and P. E. Young, "Tracking Antenna Arrays for Near-Millimeter Waves." IEEE Trans. on Antennas and Propagat., vol. 31, pp. 512-515, May 1983. 2. P. P. Tong, D. P. Neikirk, P. E. Young, W. A. Peebles, N. C. Luhmann, Jr., and D. B. Rutledge, "Imaging Polarimeter Arrays for Near-Millimeter Waves." IEEE Trans. on Microwave Theory and Techniques, vol. 32, pp. 507-512, May 1984, 4 Introduction Bibliography 1. Pp. G. Steffes and R. A. Meck, "Prototype Tests Secure Millimeter Communications." Microwave Systems News, vol. 10. pp. 59-68, October 1980. 2. G. D. O'Clock, Jr., C. Hendrickson, and W. Schaefer, "Optical and MM-Wave Links Face-Off for the Future." Microwave, vol. 21, pp. 59-63, March 1982. 3. K. J. Button and J. C. Wiltse (ed.), "Millimeter Systems." Series on Infrared and Millimeter Waves, vol. 4, Academic Press, New York, 1981, 4. J. W. Dees, G. P. Kefalas, and J. C. Wiltse, "Millimeter Wave Communications Experiments for Satellite Applications." Proceedings IEEE International Conference on Communications, San Francisco, pp. 22-20 to 22-26, June, 1970. 5. "Over $35 Million Sparking 30/20-GHz Satcon." Microwave Systems News, vol. 10, pp. 19-22, 26, Oct. 1980. 6. N. C. Luhmann, Jr., "Instrumentation and Techniques for Plasma Diagnostics: An Overview." Series on Infrared and Millimeter Waves, vol. 2, Academic Press, New York, 1979. 7. W. Kunz and G. Dodel, "On the Measurement of Poloidal Field Distributions in MTokamaks by Far- infrared Polarimetry." Plasma Phys., vol. 20, pp. 171- 174, 1978. 5 Introduction 8. J. Eberhart, "Vega and Co.: What’s Being Born Out There?" Science News 124, pp. 116, 1983. 9. J. P. Hollinger, J. E. Kenney and B. E. Troy Jr., ."A Versatile Millimeter-Wave Imaging System." IEEE Trans. Microwave Theory Tech. MTT-24, pp. 786-793, 1976. 10. D. T. Hodges, F. B. Foote, E. E. Reber and R. L. Schellenbaun, "Near-millimeter Wave Radiometric imaging." 4th Int. Conf. Infrared Millimeter Waves, IEEE Cat. No. 79CH1384-7 MTT, p. 49, 1979. 11. J. Waldman, H. R. Fetterman, P. E. Duffy, T. G. Bryant and P. E. Tannenwald, "“Submillimeter Model Measurements and Their Applications to Millimeter Radar Systems." 4th Int. Conf. Infrared Millimeter Waves, IEEE Cat. No. 79CH1384-7 MTT, p. 49, 1979. 12. J. A. Gagliano and J. J. McSheehy, "Airborne Millimeter Wave Radiometer for 94/183 GHz High Altitude Atmospheric Measurements." 6th Int. Conf. Infrared Millimeter Waves, IEEE Cat. No. 81CH1645-1 MTT, Th-4-6 1981, 13. A. R.« Gillespie and T. G. Phillips, "Array Detectors for Millimeter Line Astronomy." Astron. Astrophys. 73, p. 14, 1979, 14. K. S. Yngvesson, T. L. Korzeniowski, R. H. Matthews, P. TT. Parrish and T. C. L. G. Sollner, "Planar Millimeter Wave Antennas with Application to Monolithic Receivers." SPIE Proceedings , vol. 337, Millimeter Wave Technology, 1982. H. R. Fetterman, C. D. Parker, P. E. Tannenwald 6 Introduction and A. G. Cardiasmenos, "Printed Dipole-Schottky Diode Millimeter Wave Antenna Array." SPIE Proceedings » VOl. 337, Millimeter Wave Technology, May 1982. 16. D. P. Neikirk, D. B. Rutledge and M. S. Muha, "Par-Infrared Imaging Antenna Arrays." Appl. Phys. Lett. 40, pp. 203-205, Feb. 1982. 17. D. P. Neikirk, P. P. Tong, D. B. Rutledge, H. Park and P. E. Young, "Imaging Antenna Array at 119 um." Appl. Phys. Lett. 41, p. 329, 1982, — 18. G. R. Valenzuela, "Depolarization of EM Waves by Slightly Rough Surfaces." IEEE Trans. Antennas Propagat., vol. AP-15, pp. 552-557, July 1967. 19, I. M. Hunter, "The Application of a Two Channel Circularly Polarized Radar to the Suppression of Rain Echoes." Radar Research Establishment Journal, vol. 36, pp. 1-37, 1955. 2/LINEARLY POLARIZED ANTENNA ARRAY FOR POLARIZATION IMAGING This chapter presents an integrated-circuit antenna array that images both polarization and intensity. The array consists of a row of antennas that lean alternately left and right. creating two interlaced sub- arrays that respond to different polarizations. The arrays and the bismuth-bolometer detectors are made by a photoresist shadowing technique that requires only one photolithographic mask. The array has measured polarization at a wavelength of 800 »m with an absolute accuracy of 0.8 degrees and a relative precision of 7 arc minutes, and has demonstrated nearly diffraction- limited resolution of a 20-degree step in polarization. Applications of polarimeters The polarization of an electric field is important in many studies and applications. Natural objects de- polarize a wave while man-made objects do not. The polarization response can separate the two types of radar targets [1]. In plasma studies, both the magnetic field and electron density have to be monitored. The polarization change by Faraday rotation supplies magnetic field information [2]. The same basic idea is used in some semiconductor measurements [3]. In biochemistry. sugar concentration can be measured by the 8 Linear Antenna Array amount of rotation of the transmitted wave [4]. Some existing polarimeters from optics to microwaves A variety of different polarimeter schemes have been implemented. In microwaves, two linearly polarized antennas measuring orthogonal components of an electric field form a polarimeter [5]. In optics, the two orthogonal components can be split by a Wollaston prism and measured independently [6]. In millimeter waves, three other methods have been used for plasma diagnostics. In one, the polarization is modulated by a ferrite and one polarization component is measured after it passes through the sample material [7]. Two other schemes avoid the ferrite by frequency mixing techniques. In the first scheme, two waves with slightly different frequencies and opposite circular polarization are combined, generating a wave rotating at the beat frequency [8]. The change in phase due to the wave passing through the sample is proportional to the polarization change. This measurement is independent of fluctuations in source power. In the final scheme, two waves with slightly different frequencies but orthogonal linear polarization are generated [9]. In this last method, the amplitude of the beat signal gives the polarization changes. In all these methods, imaging can only be done by scanning or by adding more detectors. In this chapter, an integrated circuit antenna array with bolometer detectors will be demonstrated that allows polarization and intensity imaging without scanning at submillimeter wavelengths [10]. 9 Linear Antenna Array The linearly polarized antenna array Fig. 1 is a drawing of the array designed for a fused- quartz substrate. The antennas slant alternately to the left and right. Fig. 2 is a photograph of an array in its package. All the antennas are linearly polarized. The idea is that there are effectively two interlaced Subarrays (left-leaning and right-leaning) that sample two different polarization components of the image. One can recover the two components everywhere by interpolating between samples. Once the two components are known, the polarization can be calculated at all positions. Antenna spacing for diffraction-limited images The spacing between antennas should be small enough to achieve the diffraction-limited resolution of the optics. Rutledge et al. show that, in order to achieve diffraction limited resolution in intensity imaging arrays, the intervals must be no greater than agf*/2, where 4g is the dielectric wavelength and f* is the system f-number [11]. The f-number here is not the conventional f-number f/D, but is defined as 1/2sinf, where B is half the angle subtended by the exit pupil of the optical system, as seen from the image point. For heterodyne arrays, this spacing doubles. If these sampling criteria are not satisfied, aliasing problems result. In general, for the polarimeter array, the required spacing is halved because each subarray must satisfy the sampling criterion independently. It is a surprising fact, however, that the spacing requirements relax greatly if two conditions are satisfied. If the intensity varies slowly across the object and the 10 Linear Antenna Array Fig. 1. Polarimeter antenna array designed for use ona fused-quartz substrate (e, = 4). NN perecror POLARIZATION AXIS —— ANTENNA SUBSTRATE LOW FREQUENCY LEAD 11 Linear Antenna Array Fig. 2. Linearly polarized antenna array for polarization and intensity measurements in a DIP package. 05 i A 12 Linear Antenna Array polarization angles are small, then the required spacing is just rgt* even for video detection. The reason is that, under these conditions, the changes in the video detector signal become proportional to the electric ~field and the spacing requirements become those of an ordinary array for heterodyne detection. This is significant in plasma magnetic-field measurements, where the field intensity is kept relatively uniform across the object and the Faraday rotation angles are small (<10 degrees). Later, we will see this effect for a step polarization change of 20 degrees. To see this mathematically, consider an image which we characterize by an intensity I and polarization angle ¢ that are functions of the position x. The antennas in the polarimeter array receive power P given by P = AlI(x)cos?(6+¢(x)) (1) where A is the effective area for the antenna and @ is the polarization axis of the antenna. The - sign applies if the antenna leans to the left. and the + sign applies if the antenna leans tothe right. For small #¢, we may write this as I P = A(sin 26) (sana +14) (2) If we interpret the power received by each antenna as a sample of the image, and ignore the constant Asin26, there are two distinct parts of the spatial frequency spectrum of the sampled image, shown in Fig. 3. The spectrum of I/2tané is centered at f=0 and repeated at multiples of the sampling frequency f,. The spectrum of I¢, however, is displaced by f,/2. This arises from the alternating + and - sign in (2). The sampling frequency 13 Linear Antenna Array Fig. 3. The normalized filter response and sampled-image spectra for the array. F, is the sampling frequency. SPECTRUM FILTER RESPONSE OF 1/2 TAN @ SPECTRUM OF I¢ -{,/2 O f,/2 fs SPATIAL FREQUENCY f 14 Linear Antenna Array should be large enough so that these spectra do not overlap. This will allow us to recover the polarization angle ¢(x). If I is slowly varying. its spectrum will be narrow band, and the spectrum of I¢ will be only Slightly wider than that of ¢. In this case, f, must be slightly higher than twice the highest spatial frequency of ¢. When ¢ is small, it is proportional to the component of the electric field perpendicular to the ¢- axis as shown in Fig. 1. This means that the diffraction limited cut-off frequency for ¢ is the same as the cutoff frequency for fields given by 1/(224£*) [11]. The required sampling period. or antenna spacing, is then rgt*. This argument has been given for video detection, but the result is the same for heterodyne detection. To recover ¢, we filter the data to recover I and I¢ separately, and then divide the latter by I to get the polarization angle ¢ that we seek. In our measurements, where the intensity does not change in time, we normalized the response of each detector so that it was not necessary to filter for I or divide it out. The filter we used to recover ¢ was to take the difference between an antenna signal and the mean of the signals on the adjacent antennas. This gives the normalized filter response sin? (nf/f£,) shown in Fig. 3. The design and properties of the antenna array The array was designed by modeling at the microwave wavelength of 30 mn. The feed patterns for several different planes are shown in Fig. 4. Only the pattern on the dielectric side is shown. The patterns on the air side are at least 10 dB down. The antenna impedance has not been measured, but previous experiments with 15 Linear Antenna Array Fig. 4. Feed patterns measured on a scale model at 10 GHz. 90° 16 Linear Antenna Array bow-tie antennas indicate that it should be nonresonant with a resistance of 150 9. One interesting feature of the array design is the slit between adjacent antennas (see Fig. 1). Without the slit, the antennas were elliptically polarized with an axial ratio (definition at the end of the chapter) of about 3. With the slit, the axial ratio in the microwave measurement was more than 140. After the microwave tests, a scaled-down integrated circuit version was then built for a wavelength of 800 pm. A photoresist shadowing technique to make the array The array was fabricated with only one mask and one photoresist exposure by a shadowing technique. The idea was to use a 2 pm wall of photoresist to cast a shadow during the silver evaporation. The photoresist pattern had openings for both the antennas and a 3 pm wide channel connecting the antennas (Fig. 5). Chlorobenzene was used to make an overhanging lip on the photoresist to aid the lift-off process [12]. A silver layer 80 nm thick for the antennas was evaporated first at an angle of 70 degrees. The photoresist wall cast a shadow so that no silver reached the substrate in the channel. Then 200 nm of bismuth were evaporated at normal incidence. The photoresist was removed, leaving behind an antenna array with bismuth bolometers where the channels were previously. One problem arose in fabrication. Often silver strips formed along the wall of the photoresist channel. These did not break when the photoresist was removed and shorted out the bismuth bolometers. This problem was solved by making corners in the photoresist pattern to break the silver strips, so that they were removed with the photoresist (Fig. 6). Fig. 7 shows the finished device. This channel process 17 Linear Antenna Array Fig. 5. Cross-sectional view showing photoresist channel. Ea Bi ME Ag Bi EVAPORATION 0 PUOTONESIS DIRECTION CHLOROBENZENE INDUCED LIP Ag EVAPORATION DIRECTION (QUARTZ SUBSTRAT i RO i re bi 18 Linear Antenna Array Fig. 6. Top view with photoresist corners that break off silver strips. ‘ TOO ne RRR KKK ND om BR KKK oN | Bee SERRE SKN s SOLOS SOON ecaareeaareracacenseacnneeaan SKS CR 2 ' <, ODO OCLS, O SRR Peace on ones ON EDGE xXx SSN SOMO SO SSO M ESKOM KOKNO SSNS SOS MK NK OONS SOX OSS SKK SSS EN KON ONIN SRO? SSRN ERX KKRK RK SX SSG ERX oS Kesg> CHANNEL FOR SSxd, Bo Ag STRIP ox So Bi BOLOMETER SSE RSS DISCONTINUOUS > Reccuneeerereeceetaeesnens BS aT PHOTORESIST ss q SSSR RRR, RBS CORNER x SRS SSK —rapmre [SSxsxxxr SOROS XXX esenececereeen eiracetacenetaters Ag ON [XXXRKKR Me KKK KS KK PHOTO POY RRO SOCKS BKK RRR ee Bg RRR YI/ OMY PRES RS KKK % ee SO8 19 Linear Antenna Array Fig. 7. Photomicrograph with liftoff showing bismuth bolometers. The quartz substrate appears black. 20 Linear Antenna Array is simpler than the two-step process used to make the two-dimensional tracking antenna array in Chpt. 4. Also the measured electrical NEP is 5x1071%w/(Hz)71/2 at a modulation frequency of 100 kHz, which indicates that _the detectors have a higher sensitivity as well. Optical set-up The polarimeter antenna array was tested at a wavelength of 800 pm in the optical system shown in Fig. 8. The source was a Thomson-CSF carcinotron. A wire-grid polarizer ensured that the beam was linearly polarized [13]. Both the collimator lens and objective lens were made of polyethylene and had a diameter of 11.5 cm and a focal length of 12.7 cm. The beam was chopped at 100 Hz. In two experiments, a crystal-quartz half-wave plate (8.42 mm thick) was used to rotate the polarization a known amount [17]. In the third experiment the half-wave Plate was replaced by a chevron aluminum grid as an imaging target. The grid is shown in Fig. 9. The fused-quartz substrate lens had a diameter of 25 mm. The system £* was 0.8. For general polarization and intensity imaging, the spacing between the antennas for video detection should be no more than 0.2 Aq» In our experiments, however, the intensity varied slowly across the array and the polarization angles were small, so that the required spacing is four times larger, 0.8 rq Our array Spacing, 0.5 4g, satisfied this more relaxed criterion. The signal captured by the antenna and detector was measured by a model 5204 PAR lock-in. About 5 mW power reached the objective lens. The system responsivity was 0.5 V/W for a 90 © bolometer biased at 150 mV. 21 Linear Antenna Array Pig. 8 Linear polarized antenna array optical set-up. AS ats SUBSTRATE WITH ARRAY ANTENNA ON BACKSIDE vache LENS { I CHEVR WAVE WIRE GRID WIRE SRD PLATE =e POLARIZER AS OBJECT rs cuokeer coutlvsron LENS CARCINOTRON OUTPUT FROM OPEN-ENDED WAVEGUIDE 22 Linear Antenna Array Fig. 9. Photomicrograph of the chevron grid. The light strips are aluminum, 20 pm wide, spaced 20 pm apart, and 300 nm thick. The fused quartz substrate appears black. The grid is made by standard contact photolithography -and liftoff. 23 Linear Antenna Array Measuring the polarization angle -Two experiments would be shown here, the first one verified that the antennas were linearly polarized and the second one tested out the response of the array to small angular changes in polarization. From the measurements, the polarization axis was found to be 38 degrees to the left from the ¢-axis as shown in Fig. 1 for the left-leaning antenna and 38 degrees to the right for the right-leaning antenna. Fig. 10 shows the response of a single antenna to different polarization angles. Comparison with the theoretical cosine-squared curve shows that the antennas are linearly polarized. In the next measurement, the polarization across the object plane was constant, and could be varied by rotating the half-wave plate. Fig. 11(a) shows the measured polarization angle from the array for one setting of the half-wave plate. The standard deviation of the data is 0.8 degree. The change in the measured polarization angle when the half-wave plate was rotated is shown in Fig. 11(b). Here the standard deviation among the data in measuring a 6 degrees change is 5 arc Min, and their average value differs from the theoretical value by 7 arc minutes. Imaging a step change in polarization The next experiment intended to prove that the array gave diffraction limited images. But before presenting that, this section will first illustrate the theoretical diffraction limited optical response of a step change in polarization. A step change in the polarization of the electric field is generated by a chevron wire grid 24 Linear Antenna Array Fig. 10. Single antenna response as a function of the polarization angle. | | T T THEORY re) = e EXPERIMENT _ 120 tele ANTENNA RESPONSE, pV O @ O (e) ff O ! O ! { 0° 40° 80° I20° I60° RELATIVE POLARIZATION ANGLE 25 Linear Antenna Array Fig. 11. The array response to small polarization angle. (a) The measured polarization angle for one setting of the half-wave plate. (b) The change in the measured polarization angle caused by rotating the half-wave . plate. { aa } i a q 15eh 4 liu a) 10° ~ Zz x x Tt oseh 4 6 x F oe? -------------~---- x---------- 4 N ° fad -0.5 a < THEORY ---- 5 -1.0°R x EXPERIMENT x J “15°h * 4 { ij } } j } 04 -O2 re) 0.2 0.4 0.6 DETECTOR POSITIONS IN THE IMAGE PLANE, mm 1 7 T ] We Se Se on = =—— zZ <q z g aL _ O ZN as 9) ; Fe < eT * ~ ~ cay - ~K 9 THEORY 2 EXPERIMENT 2 * éL_t { ! ° -0:4. 0,2 a) 0.2 0.4 0.6 DETECTOR POSITIONS IN THE IMAGE PLANE, mm 26 Linear Antenna Array polarizer [13]. Fig. 9 is a photograph of the grid. The frequency of the wave is 375 GHz. The electric field changes from leaning to the left 10 degrees to leaning to the right 10 degrees across the chevron wire grid . polarizer. Fig. 12 shows the co-ordinate system. The electric field just passes through the grid can approximately be resolved into two components: 1 x>0 i: sgn(x) = 0 for x=0 -1 x0 jo: 1/(tan 10°). At the image plane the jth component of the polarization of the electric field can be found by Fourier methods [14] to be: a w t F oo . . 2 pexFz 1 me —jaxfz tafe — sint t: I, [/= san(z)e dz| ell? df = - | (*) dt 2 G- == S1(2nF 2) with F electric field cut-off frequency of the optical system. 1/(2r5£*) wheredg= the dielectric wavelength of the medium in which the image is formed fH = system f£f number. [11] Si as the Sine integral. The jth component does not change since it is constant across the whole field of view of the optical system. 27 Linear Antenna Array Fig. 12. The co-ordinate system for the wire grid. ELECTRIC FIELD wv GRID See 28 Linear Antenna Array The polarization of the electric field as a function of x can be found by combining the ith and the jth components. The results are drawn up in Fig. 13 for the optical system given above. The antenna array imaged the chevron aluminum grid. After the wave passes through the grid, the electric field is perpendicular to the metal lines. This causes a 20 degree step change in polarization at the center of the chevrons. Fig. 13 shows the polarization image, along with the images predicted by geometrical optics (a step change) and by diffraction theory for diffraction limited optics, z ¢(z) = arctan [25s (5) tan) (3) where Si is the sine integral [15] and @, is the step change (20 degrees in this case). The agreement between the experimental points and the diffraction theory indicates that the array is nearly diffraction-limited. This demonstrates that, for slowly varying intensities and small angles, diffraction limited resolution may be achieved by sampling much less often than the general sampling criterion. Fig. 13. POLARIZATION ANGLE, DEGREES 29 Linear Antenna Array Polarization image of the chevron grid. POSITION ACROSS THE OBJECT PLANE ,mm “6 -4 -2 0 2 4 6 a O T q I q 1 ! x THEORY, GEOMETRICAL -----] OPTICS THEORY, DIFFRACTION —— LIMITED OPTICS EXPERIMENTAL x ! ~O6 -04 “OB 0 0.2 04 0.6 DETECTOR POSITION IN THE IMAGE PLANE, mm 30 Linear Antenna Array Appendix : common definitions in antenna polarization Four different terms commonly used will be discussed. They are “1, Axial ratio 2. Ellipticity 3. Eccentricity 4. Cross polarization ratio. Any polarized electric field can be represented as E = x+ betTy Or aS a matrix _ [seir4 . The two major axes of an ellipse are related to the above terms by a*=cos7yt5"sin7y+8sin2yvcosy b7=sin7y+8*cos2y-8sin2ycosy where wv is related to the wave as (1-87) sin2y = 2$cosycos2y. All these terms are shown in Fig. 14 [16]. 31 Linear Antenna Array Fig. 14. A right-handed circularly polarized (traditional definition) ellipse. The electric field is given by E = x- iy Ay xV 32 Linear Antenna Array Now the different representations for an elliptical wave Or antenna are: _Axial ratio (Semi-major axis)/(semi-minor axis) = a/b = 1/z;tan2 where sin29 = (sin2a)siny and tane = 8. It may also be defined as whereE, =right-handcircularly polarized component of the wave; E,, = left-handcircularly polarized component of the wave. Fig. 15 shows the axial ratio of some types of ellipse. Ellipticity (Semi-minor axis)/(semi-major axis) = b/a =z;tan® Eccentricity (1-(b2/a2))1/2 Cross polarization ratio of an antenna 1010g;9(Max. power received/Min. power received) One way to get the maximum and minimum power received by 33 Linear Antenna Array Fig. 15. The axial ratio of a few types of ellipse. AXIAL RATIO x A y -l a) _, ( | AC oe D 34 Linear Antenna Array the antenna is by rotating a linearly polarized antenna 180 degrees and monitoring the response of the antenna. The relationship between the axial ratio and the cross polarization ratio is : Cross polarization ratio = 20Log,9(Axial ratio). 35 Linear Antenna Array Bibliography 1. G. R. Valenzuela, "Depolarization of EM waves by Slightly Rough Surfaces." IEEE Trans. Antennas Propagat., vol. AP-15, pp. 552-557, July 1967. 2. N. C. Luhmann, JdJr., "Instrumentation and Techniques for Plasma Diagnostics: An Overview." Series on Infrared and Millimeter waves, vol. 2, Academic Press, New York, 1979, 3. E. K. Galanov, G. N. Potikhonov, V. V. Sorokhin, and O. M. Sheshin, "Infrared Polarimeters and Questions of Method Relating to Contactless Measurement of Free-~carrier Concentrations in Semiconductors." Ind. Lab., vol. 42, pp. 1530-1533, Oct. 1976, 4. J. W. Gates, "An Automatic Recording Saccharimeter." Chem. Ind., pp. 190-193, Feb. 1958. 5. S. Suzuki and A. Tsuchiya, "A Time-sharing Polarimeter at 200 MC." IRE Proc., vol. 46, pp. 190- 194, Jan. 1958. 6. E. Hecht and A. Zajac, Optics, 4th edition. Addison-Wesley. Feb. 1979, p. 238. 7. C. H. Ma, D. P. Hutchinson and K. L. Vander Sluis "A Modulated Submillimeter-laser Polarimeter for the Measurement of the Faraday Rotation by a Plasma." Appl. Phys. Lett., 34, pp. 218-220, Feb. 1979, 8. G. Dodel and W. Kunz, "A Far-infrared 'Polari- interferometer’ for Simultaneous Electron Density and Magnetic Field Measurements in Plasmas." Infr. Phys., vol. 18, pp. 773-776, 1978. 36 Linear Antenna Array 9. H. Soltwisch and T. F. R. Equipe, "Experimental Test of Far-infrared Polarimetry for Faraday Rotation Measurements on the TFR 600 Tokamak." Infr. Phys. vol. 21, pp. 287-298, 1981, 10. P. P. Tong, D. P. Neikirk, P. E. Young, W. A. Peebles, N. C. Luhmann, Jr. and D. B. Rutledge, "Imaging Polarimeter Arrays for Near-Millimeter Waves." IEEE Trans. on Microwave Theory and Tech., vol 32, pp. 507- 512, May 1984, 11. D. B. Rutledge, D. P. Neikirk, and D. P. Kasilingam, "Integrated Circuit Antennas." Series on Infrared and Millimeter Waves, vol. 10, Academic Press, New York. 12. M. Hatzakis, B. J. Canavello, and J. M. Shaw, "Single-step Optical Lift-off Process." IBM J. Res. Develop., vol. 24, pp. 452-460, July 1980. 13. G. R. Bird and M. Parrish, Jr., "The Wire Grid as a Near-infrared Polarizer." J. Opt. Soc. Am., vol. 50, 14. J. W. Goodman, Introduction to Fourier Optics. New York: McGraw-Hill, Chpt. 6, 1968. 15. M. Abramowitz and I. A. Stegun, Handbook of Mathematical Functions. Dover Publication, New York, Chpt. 5, 1965. 16. M. Born and E. Wolf, Principles of Optics, 6th edition. Pergamon Press, Chpt. 1. 17. F. A. Jenkins and H. E. White, Fundamentals of Optics, 4th edition. New York, McGraw-Hill, 1976, p. 567. _3/CIRCULARLY POLARIZED SPIRAL ANTENNA ARRAY This chapter discusses the performance of a circularly polarized antenna on a dielectric. It is an equiangular spiral antenna. The axial ratio, antenna patterns and efficiency of the antenna will be presented. Also a new antenna pattern measurement technique in the millimeter waves for integrated-circuit antennas will be shown. Circularly polarized spiral antenna : motivation Circularly polarized antennas are attractive for many applications. Because of their spherical shape, their reflections reverse the sense of an incident circular polarized wave, and so a circular-polarized radar can suppress much rain clutter [1]. Also they give identical response to a linearly polarized wave whose electric field is at any orientation. Fig.1 shows the circularly polarized spiral antenna on quartz. Its E and H plane antenna patterns have 3 dB beamwidths less than 12 degrees. Measurements indicate that the efficiency is quite high, 86%. This is much better than the performance previously reported for bow- tie, which is around 25%. 38 Circular Antenna Array Fig.1 Circularly polarized antenna array in a DIP package. 39 Circular Antenna Array The equiangular spiral antenna in free space The equiangular spiral antenna in free space has previously been studied [2]. The antenna can be described in polar co-ordinates by the equation, r = keal¢-a), (1) It is called equiangular because the angle between the Slope of the curve and the radius is always a constant. It is given by, cos@ = a/(a2+1)1/2 where 6 is as shown in Fig. 2. It is a frequency-independent antenna since all shapes are described in terms of angles [3]. Bandwidth in excess of 20 to 1 has been achieved. The design and impedance of the spiral on dielectric The integrated-circuit spiral antenna on quartz has the Same shape as the free space spiral. A lens was placed on the back of the substrate. This substrate lens takes advantage of the fact that antennas are most sensitive to radiation from the substrate side [4]. Pig. 3 shows the design of the antenna on a substrate with a Gielectric constant of 4. It has been found experimentally that the antenna will be circularly polarized (axial ratio < 1.2) if the detector length is less than 0.04 4g and the antenna is more than 1.25 ig in diameter. Ag is the dielectric wavelength. Fig. 3 shows a photograph of a few of the antennas in an integrated-circuitpackage. 40 Circular Antenna Array Fig. 2. An equiangular spiral. a(d-a) r=ke 41 Circular Antenna Array Fig. 3. Circularly polarized equiangular spiral antenna designed for use on a fused-quartz substrate (e, = 4). A 1,25 5 0,04 4 —+ k— eal $-31r/2) ea ($-77) ef > e! ($-71/2) Detector Antenna Low Frequency Lead 42 Circular Antenna Array The distance between the antennas is 0.42 Aa The antenna is self-complementary [5]. We can estimate its impedance as [6], zg? = n7/4 where ny = No/ (et /2 with ny = the impedance of the free space, and « = mean of the two dielectric constants. With fused quartz as the substrate, the terminal impedance is close to 1208. Bi bolometers serve as the detectors. The leads extending from the edge of the spiral are for biasing and low-frequency detection. Integrated-circuit fabrication of the spiral The spiral antenna is made by integrated-circuit techniques. An HP plotter drew the mask for the spiral. The integrated-circuit was then made by lift-off techniques. The antennas are made of silver two thousand angstroms thick. A chlorobenzene process [7] improved the lift-off edges. Two thousand angstroms of Bismuth were then evaporated at a 45-degree angle to help solve the step coverage problem [8]. Fig. 4 is a photomicrograph of the antenna with a Bi bolometer at the middle. The resistance of the bolometer is about 170 Q, Three characteristics of the antenna -- antenna axial ratio, antenna pattern and the efficiency of the spiral will be presented. 43 Circular Antenna Array Fig. 4. Photomicrograph of one spiral antenna. The bright area is the silver antenna while the dark area is the quartz substrate. The dark line in the middle is a . Bi bolometer. 44 Circular Antenna Array The axial ratio of the spiral on dielectric. Axial ratio of an ellipse is defined as (Semi-major axis)/(semi-minor axis). It may also be defined as where Ep = right-hand circularly polarized component of the wave. E,, = left-hand circularly polarized component of the wave. Fig. 5 shows the optical set-up to measure the antenna axial ratio in 94 GHz. The source was a linearly polarized radiation at 94 GHz. The wave was electronically modulated at 1kHz. It was collimated and passed through a quarter-wave plate and an objective lens and received by the spiral antenna at the back of a substrate lens. The objective lens is made of polyethylene with a 15-inch focal length and a 6-inch diameter. The substrate lens is aplanatic [9], and is made of fused quartz with a 1.97-inch radius. The quarter-wave plate is of crystal quartz [10]. Depending on its orientation, it changes a linearly polarized wave from linear to elliptical to circular wave. The theoretical response of a circularly polarized antenna to the above set-upis P, = 1-sin26 where 6 is the angle between the optical axis and the direction perpendicular to the linearly polarized wave as shown in Fig. 6 and 45 Circular Antenna Array Fig. 5. The optical set-up to test the response of the antenna to an elliptically polarized wave. 46 Circular Antenna Array Fig. 6.The relationship between the incident electric field direction and the orientation of the uniaxial crystal. UNIAXIAL CRYSTAL INCIDENT ELECTRIC FIELD LEPINE LLNS ISI EOE i i . L : : 2 | 47 Circular Antenna Array P. = power received by the antenna. r Fig. 7 shows the theoretical and experimental results. The axial ratio of the antenna is less than 1.2. Antenna pattern measurements at millimeter wavelengths A quasi-optical system is used to measure the antenna pattern of an integrated-circuit antenna on a dielectric. The dielectric antenna is positioned at the center of the radius of curvature of a hemisphere, so that any rays reaching the antenna are not refracted by the hemispherical lens. By rotating the lens about its center of the radius of curvature, one can measure the antenna response to different directions. This will then give the antenna pattern. Fig. 8 shows the idea. There are a few things that one has to pay attention to in pattern meaSurements. First, the set-up has to be in an anechoic chamber as shown in Fig. 9. Absorbers (supplied by Plessey) are needed at the back of the hemisphere as well. The antenna should be carefully centered. Fig. 10 is a diagram of the optical set-up together with the data-collecting instruments to measure the patterns of the antennas in the millimeter wavelengths. A klystron transmitted a 94 GHz wave through a 24 dB gain horn. The horn had a1 inch by 1 inch mouth. The wave was electronically modulated at 1 kHz. A hemispherical lens was placed 30 inches away from the mouth of the horn. The lens was made of fused quartz with a 1.97 inch radius. Fig. 11(a) is a picture of the lens in its holder with two stepper motors, one on the side and the other at the bottom. The motor can give 4 steps per 48 Circular Antenna Array Fig. 7. The theoretical and experimental response of the antenna to elliptically polarized wave. 0.5- — THEORY @ EXPERIMENT ANTENNA RECEIVED POWER (RELATIVE) INCIDENT WAVE AXIAL RATIO 49 Circular Antenna Array Fig. 8. Antenna pattern measurement idea. . ANTENNA gpa AT THE CENTER a SOURCE HEMISPHERE 50 Circular Antenna Array Fig. 9. Photograph of the antenna pattern measurement optical set-up in an anechoic chamber. 51 Circular Antenna Array Fig. 10. Diagram of the optical set-up together with the data-collecting instruments. ROTATIONAL ANTENNAS STAGE WITH —_ LENS WITH morors HORN STEPPER \ DETECTORS SUPPLY MOTORS ' i) ‘N XN 1 | ‘ ] ! \ | \ \ i . MOTORS ‘. | CONTROLLER | | ‘. ot COMPUTER ATOD CONVERTER 1AS SUPPLY - 94 GHz SOURCE AND CONTROL[-] AMPLIFIER CIRCUITRY 52 Circular Antenna Array degree. Each motor needs a 28 V and 2 A power supply. The lens could be rotated by the stepper motors about its center of curvature in both the @ and ¢ directions _in the spherical co-ordinates. The stepper motors were controlled by an IBM personal computer through a Tecmar PC-Mate Stepper Motor Controller. Fig. 11(b) shows the lens with the antennas. As the lens rotated, the antenna would couple in different power corresponding to its pattern. The power was amplified by a Princeton Applied Research Model 124A Lock-In Amplifier with a model 118 Differential Preamplifier. Then the signal was fed to the same personal computer through a Data Translation Model 2801 analog to digital converter. Data were stored and manipulated as desired. The program to rotate the stepper motors and then take the data is as shown in Appendix 3. The measurement methods and set-up were tested out by measuring the antenna patterns of a long wire antenna on a dielectric. It was described in Appendix 2. The results give us confidence in the measured patterns. The antenna patterns of the spiral on dielectric Fig. 12 gives the two principal patterns of the spiral antenna. Due to symmetry of the spiral, the two patterns should be identical. Both of them had a center peak. The E-plane 3 dB beamwidth was 11.5 degrees, while the H-plane 3 dB beamwidth was 8.5 degrees. In comparison, the narrowest 3 dB beamwidth of the modified bow tie array in Chapter 2 was 67 degrees and the two- dimensional array in Chapter 4 was 34 degrees. 53 Circular Antenna Array Fig. 11(a). Hemispherical lens in its holder showing the two stepper motors. 54 Circular Antenna Array Fig. 11(b). Lens with the antennas. 55 Circular Antenna Array Fig. 12. The antenna patterns of the spiral antenna. Experimental points were taken every 2 degrees. { q { { | | | l 7 —— E-PLANE = ~---H-PLANE _ va OW 7 Ww > 90° Ze “%S Wd <t O.5- Ye 4 O _J fe Ld i Co T “ < L Z = Zz L < a Z = _ 0 0 20 40 60 80 DEGREES 56 Circular Antenna Array The efficiency of the spiral on dielectric The optical set-up to measure the antenna efficiency is the same as that shown in Fig. 5, except that the _collimator lens and the quarter-wave plate were removed. In order to ensure almost plane wave incidence onto the antenna, the distance between the objective lens and the horn was [11] R> 2*D7/2 where D is the diameter of the lens and 4 is the free space wavelength. The objective lens was a TPX [12] plano-convex lens 4.8 cm in diameter and 4.8 cm in focal length. The antenna was at the back side of the substrate lens. The substrate lens was an aplanatic lens. It was made of Dynasil 4000 fused quartz with a 1/2 inch radius. The input power of the antenna was assumed to be all the power incident on the objective lens. About 43% of the power was captured by the center antenna in Fig. 3 after focusing the optics, and no power registered in the two adjacent antennas. Since the antenna was circularly polarized and the incident wave was linearly polarized, the antenna could only capture half the incident power. Hence, if the incident wave was circularly polarized, the antenna efficiency should be 86%. 57 Circular Antenna Array Bibliography 1. I. M. Hunter, "The Application of a Two Channel Circularly Polarized Radar to the Suppression of Rain Echoes." Radar Research Establishment Journal, vol. 36, pp. 1-37, 1955. 2. J. D. Dyson, "The Equiangular Spiral Antenna." IRE Trans. Antennas Propagat., 7, pp. 181-187, 1959, 3. V. He. Rumsey, "Frequency Independent Antennas." IRE Nat’l Convention Record, Pt. I, pp. 114-118, 1957. 4. D. B. Rutledge and M. S. Muha, "Imaging antenna arrays." IEEE Trans Antennas Propagat., AP-30, pp. 535-540, 1982. 5. J. D. Kraus and K. R. Carver, Electromagnetics, 2nd edition . New York, McGraw-Hill, Chpt. 14. 6. D. B. Rutledge, D. P. Neikirk, and D. P. Kasilingam, "Integrated Circuit Antennas". Series on Infrared and Millimeter Waves, vol. 10, Academic Press, New York. 7. M. Hatzakis, B. J. Canavello, and J. M. Shaw, "Single-step Optical Lift-off Process." IBM J. Res. Develop., vol. 24, pp. 452-460, July 1980. 8. P. P. Tong, D. P. Neikirk, P. E. Young, W. A. Peebles, N.C. Luhmann,Jr., D. B. Rutledge, "Imaging polarimeter arrays for near-millimeter waves." IEEE Trans. Microwave Theory Tech., vol. MTT-32, pp. 507- 512, May 1984, 58 Circular Antenna Array 9. F. A. Jenkins and H. E. White, Fundamentals of Optics, 4th edition. New York, McGraw-Hill, 1976, Chpt. 9. . 10. Ibid. Chpt. 27. 11. H. Jasik, Antenna Engineering Handbook, ist edition, McGraw-Hill Book Company, 1961, Chpt. 34. 12. G. W. Chantry, J. W. Fleming, P. M. Smith, M. Cudby and #H. A. Willis, "Far Infrared and Millimeter-wave Absorption Spectra of Some Low-loss Polymers." Chen. Phys. Let., vol. 10, no. 4, pp. 473-477, August 1971. 4/TWO-DIMENSIONAL TRACKING ANTENNA ARRAY This chapter describes a two-dimensional monolithic array that gives the elevation and azimuth of point source targets. The array is an arrangement of rows and columns of antennas and bismuth bolometer detectors on a fused quartz substrate. Energy is focused onto the array through a lens placed on the back side of the substrate. At 1.38 mm with a 50 mm diameter objective lens, the array has demonstrated a positioning accuracy of 26 arcminutes. In a differential mode this precision improves to 9 arcseconds, limited by the mechanics of the rotating stage. This tracking could be automated to a fast two-step procedure where a source is first located to the nearest row and column, and then precisely located by scanning. With signal processing the array should be able to track multiple sources. The idea of the array Tracking millimeter wave sources without scanning were done by a two dimensional monolithic array [1]. Figs. 1 and 2 show the array designed for wavelength of 1.2mm. Because of the series connection, the signal measured on the low-frequency leads gives the total power received along a particular row or column. The strongest column Signal gives the source azimuth, and the strongest row signal gives the source elevation. More precise 60 2-D Antenna Array Fig. 1. Tracking array. (a) View from the air side Showing standard DIP package. 61 2-D Antenna Array Fig. 1. Tracking array (b) View through substrate lens. 62 2-D Antenna Array Fig. 2. Micrograph of a tracking array designed for 1.2 mm. The antennas are made of silver, and appear dark. The distance between adjacent rows (or columns) is 268 nm (7/1644) and the distance between adjacent detectors -in a single row or column is 535 pm (7/844). At its widest, the silver is 80 nm wide (2/1543). Overlapping rows and columns are insulated from each other by a layer of polyimide. | | Bi Bolometer Detector Columns \ | | i | Square Insulating s Layer | Detector | Rows { 63 2-D Antenna Array tracking information can be obtained by taking the difference between adjacent row and column signals in a way similar to optical centroid trackers [2] and microwave monopulse receivers [3]. In order to get this “improvement, however, the array must be wobbled slightly to null the difference signal. This array can locate a single source, and the question that naturally arises is: What happens when there are more point sources? If the sources all have different intensities, azimuths, and elevations, the array could track them simultaneously by identifying the row and column signals with equal magnitudes. If intensities, azimuths, or elevations are equal, there will be ambiguities. In a dynamic situation, these ambiguities INMay Only persist for short periods of time. and with appropriate processing, multiple-target tracking should be feasible. It should also be possible to follow a Single source when other sources pass through in the field of view of the array. There is an interesting connection between the tracking array and X-ray transaxial tomographic imaging. The information produced by the array is the projections of the image along two perpendicular axes. The ambiguities that arise from sources of equal intensities, azimuths, and elevations could be resolved in principle by rotating the array about the optical axis to produce different projections. In some applications, such as missile guidance, the platform may rotate anyway. In principle, an arbitrary image can be reconstructed from its projections, similar to the way it is done in X-ray transaxial tomographic imaging [4]. 64 2-D Antenna Array The design, impedance and antenna patterns of the array The array was designed by building large-scale microwave models. The model antennas were made of copper foil and the substrate was artificial dielectric with a dielectric constant of four to match the fused quartz substrates used at the millimeter wavelengths. The antennas were isolated from each other at low frequencies by masking tape. The spacing between rows and columns must be small enough so that the resolution does not degrade. A A’Ag/2 spacing is adequate [5]. The dimensions were chosen by trial and error to give adequate impedance and radiation patterns. The impedance of the model indicates the coupling efficiency for different kinds of detectors and the available bandwidth. Fig. 3 shows the model embedding impedance in ohms from 2 to 4 GHz. At the design frequency, 3 GHz. the impedance was 150 + j20 & The resonance is weak, and this should be an adequate impedance for matching to either Schottky diodes or bolometers over a large bandwidth. The radiation patterns of the array indicate how efficiently energy will be coupled into the array and also help determine the optical system. These patterns are needed because it is difficult to make accurate coupling efficiency measurements at millimeter wavelengths. Fig. 4 shows the radiation patterns into the substrate measured at a single detector in the array. The pattern for radiation into the air is at least 10 dB below the peak substrate power and is omitted. The patterns show a symmetrical beam with a half-power beamwidth of 30 degrees and the E-plane pattern shows large sidelobes that could not be eliminated. 65 2-D Antenna Array Fig. 3. The impedance of the large-scale model of the antenna. It was plotted from 2 to 4 GHz ona Smith chart. lOO I40 200 66 2-D Antenna Array Fig. 4. Antenna patterns measured at a single detector diode at 10 GHz ona scale model. The plot is linear in power. . —-— 45° --—— E-plane —— H-plane 90 in 90° 67 2-D Antenna Array Integrated-circuit fabrication of the array The array is made by conventional contact lithography and lift-off. The silver antennas were 80 nm thick. A half-micron polyimide insulating layer, Dupont Pyrallin 2555, was spun on between successive silver evaporations. Then 300 nm Bi bolometers were deposited. The particular array used in testing had bolometer resistances in the range 15 to 25 a MThis is a poor match to the antenna but the resulting signal levels were sufficient to demonstrate the array properties. After fabrication, the array was mounted in a standard dual in-line package (DIP). The full array is 32 by 32, and this size can be adjusted during wire bonding. In our experiment the arrays were 7 by 7. The number of antennas and detectors in a row or column affects the array sensitivity. This arrangement, where the output of a group antennas and detectors is combined, is called a multimode antenna [6]. A general property of multimode antennas is that the sensitivity degrades as the number of detectors increases. For the series-connected rows and columns, this is easy to understand because the total noise of a row or column is larger than the noise of a single detector. In terms of the noise equivalent power (NEP), for a row or column of n detectors, it is the square root of n times of the NEP of a single device. The NEP of the bolometers fabricated was about 107 9w/(Hz)71/2, Although an order of magnitude improvements in sensitivity can be achieved by the fabrication method shown in Chapter 2, the above sensitivity was already good enough in testing antennas and arrays. 68 2-D Antenna Array The tracking of a target Fig. 5 shows the optical set-up where the array was tested out in the submillimeter wavelength. The fused .quartz substrate lens had a radius of curvature of 6.62 mm. It was an aplanatic lens [7]. The TPX [8] objective lens had a 70 mm focal length and a 50 mm diameter. Fig. 6 shows the system patterns at 1.22 mm. The H- Plane pattern is generated when the focused spot scans across a row. This pattern is sharply peaked, resembling the Airy pattern of the optical system [9]. The E-plane pattern results when the focused spot scans along a row from detector to detector. Ideally this signal should be constant, but Fig. 6 shows some roll- off, probably resulting from lens aberrations and differences in bolometers. The next measurement demonstrated the tracking ability of the array. The source, a klystron at 1.38 mm, was translated in a plane 70 cm from the array. The power incident on the objective lens was 0.4 mW, and the output signal maxima were about 30 »V. The positions of the maxima were determined by standard sincfunction interpolation of sampling theory [10]. Fig. 7 compares the measured positions in the image plane with the predictions of geometrical optics. The average error in the image plane was 160 um, and the corresponding error in the far field was 26 arcminutes. This is five times smaller than the diffraction-limited Rayleigh resolution limit for a pair of point sources. Much more precise tracking information can be obtained by taking the difference between adjacent row or column signals. As a function of angle, this pattern (Fig. 8) is the difference between two overlapping H-plane system patterns. It was measured with the differential inputs 69 2-D Antenna Array Fig. 5. Substrate-lens coupled optical system. ~~ Antenna Array —_———n ~N TN Le L | N Dubstrate Substrate Objective Object Lens Plane 70 2-D Antenna Array Pig. 6. Near-millimeter system patterns. The source was a CO,~pumped Cy3 methyl-fluoride laser. 1.0 T T T T Measured H , plane __. Measured E ES O.8F plane \. - OF ou. . Theoretical ,f: ‘. © airy pattern, : ng / : ~~. _ O 0.6 a — & 7 @ 7 a) OD O.4F i @ IN E Yee 0.2 — _ ro) =a 0.0 pert reg ede, e *. ] rT * eecene , eee -3 -2 a | O | 2 3 Angle, Degree 71 2-D Antenna Array Fig. 7. Tracking demonstration. The grid shows the locations of the detector rows and columns. Detector Columns Q x x netector Ae Rows \ Ty x $ * —268um —> <— x Calculated Position © Measured Position 12 2-D Antenna Array of a PAR-124 lock-in amplifier. The actual tracking procedure is similar to that of monopulse radars and optical centroid trackers where the system must be oriented to null the difference pattern. We were able _to orient the system toward the source with an accuracy of 9 arcseconds, as indicated by the resulting difference signal. This accuracy was limited by the mechanics of the rotating stage rather than by the system signal-to noise ratio. 73 2-D Antenna Array Fig. 8. Magnitude of the difference pattern of adjacent rows (000), along with the two overlapping H-plane system patterns (xxx and eee). The difference signal _changes sign on each side of the null. 1.0 r) 0.8 0.6 0.4 \ H | I \, \, a 4 l Normalized Received Power -3 -2 -| 0° | 2 Angle , Degrees 14 2-D Antenna Array Bibliography -1. BP. P. Tong, D. P. Neikirk, D. Psaltis, D. B. Rutledge, K. Wagner, P. E. Young, "Tracking Antenna Arrays for Near-Millimeter Waves." IEEE Trans. Antennas and Propag., vol. 31, May 1983, 2. C. M. Crocker,Jr., "Wide Dynamic Range Instrument for Measurement of Laser Spot Position on Stationary and Moving Targets." Proc. Soc. Photo-Opt. Instrum. Eng., vol. 255, pp. 69-72, 1980. 3. M. I. Skolnik, Radar Handbook. New York: McGraw- Hill, Chpt. 21, 1970. 4. H. Barrett and W. Swindell, "Analog Reconstruction Methods for Transaxial Tomography." Proc. IEEE, vol. 65, p. 89, 1977. 5. D. B. Rutledge, D. P. Neikirk, and D. P. Kasilinganm, "Integrated Circuit Antennas." Series on Infrared and Millimeter Waves, vol. 10, Academic Press, New York. 6. D. B. Rutledge and S. E. Schwarz, "Planar Multimode Detector Arrays for Infrared and Millimeter-Wave Applications." IEEE J. Quantum Electron.., vol. 17, pp. 407-414, 1981. 7. F. A. Jenkins and H. E. White, Fundamentals of Optics, 4th edition. New York, McGraw-Hill, 1976, p. 567. 8. G. W. Chantry, J. W. Fleming, P. M. Smith, M. Cudby and H. A. Willis, "Par-infrared and Millimeter-wave 75 2-D Antenna Array Absorption Spectra of Some Low-loss Polymers." Chem. Phys. Let., vol. 10, no. 4, pp. 473-477, August 1971. 9. J. W. Goodman, Introduction to Fourier Optics. New York: McGraw-Hill, p. 64, 1968. 10. Ibid., pp. 21-25. _5/FUTURE WORK IN MILLIMETER WAVES This thesis has presented the first linearly and circularly polarized antennas on dielectrics at millimeter wavelengths. The circularly polarized antennas can be put into arrays for polarization and intensity imaging. The idea will be similar to the linearly polarized antenna arrays except that now the antennas are right-handed and left-handed circularly polarized. Also the low-frequency lead can be removed without affecting the performance of the antenna. This is because, if the high frequency does propagate into the two vertical low frequency lines, then the antenna will not be circular. Biasing and the low-frequency power detection can be done directly at the end of the spiral. This will make the antenna more compact and will allow two-dimensional imaging. Another interesting area will be combining both the linearly and circularly polarized antennas in an array. If the array works, then, through power detection alone, the array could detect the components of the electric field and the phase between them. That means there is no need to detect the phase between the electric fields directly. That simplifies the measurement procedures and set-up. This thesis has also presented the first two-dimensional 17 Future Work tracking antenna array. This array images point targets. For general targets, it would be best if output from each individual detector can be extracted. Now we are limited by the amount of electrical wires that can go to -individual detectors. The scheme to get around using wires would be to have a 2-dimensional array of spiral antennas without low frequency leads, all of them are separated from one another. Thermocouples are the detectors. Signals can be coupled out by two different methods. One is to integrate amplifiers and light emitting diodes to the detectors. Then, instead of electrical outputs, one would have optical output. Another method would be to bring an array of probes in from the back side onto the array, a separate probe for each of the spiral end to couple the signal out. A general two-dimensional imager will be possible by either of the above methods. Also of interest will be integrating sources such as IMPATT or GUNN diodes instead of detectors into the arrays of integrated circuit antennas. We have developed some very low-impedance narrow-band antennas for impedance matching in this application. _APP1/THE RESPONSE OF ANY POLARIZED ANTENNA TO ANY POLARIZED WAVE This appendix addresses the response of any polarized antenna to any polarized wave. The experimental work was done at millimeter wavelengths. Modified bow-tie antenna is the linearly polarized antenna; equiangular spiral antennas are the elliptically and circularly polarized antennas. The experimental results match with theory. The linearly, elliptically and circularly polarized antennas Linearly, elliptically and circularly polarized antennas are used. They are integrated-circuit antennas on a dielectric material. Modified bow-tie antennas are the linearly polarized antennas [1]. Fig. 1 is a drawing of a few of them designed for a fused-quartz substrate. Bismuth bolometers are the detectors. Fig. 2 is a photograph of an array of themina package. The array was designed at 375 GHz in frequency or 0.8 mm in wavelength. All the antennas are linearly polarizea with an axial ratio (definition in the Chapter 2) more than 140. Chapter 2 gives a more elaborate explanation of the antenna. Equiangular spiral antennas [2] are the 719 Polarized Antenna’s Response Fig. 1. Linearly polarized modified bow-tie antennas designed for use on a fused-quartz substrate (e, = 4). N_petector POLARIZATION AXIS ——| ANTENNA SUBSTRATE LOW FREQUENCY LEAD 80 Polarized Antenna’s Response Fig. 2. Linearly polarized antenna array in a standard DIP package. 81 Polarized Antenna’s Response elliptically and circularly polarized antennas. Fig. 3 is a drawing of a circularly polarized antenna. Itisa self-complementary antenna [3]. Again Bi bolometer is the detector. Fig. 4 is a photograph of a few of the -antennas in a package. Two are right-hand circularly polarized and the center one is left-hand circular. It was designed at 94 GHz or 3.2 mm with an axial ratio less than 1.2. Chapter 3 discusses this antenna further. The antenna becomes elliptical when there is insufficient spiral length or the detector position occupies a relatively large area. Fig. 5 shows the photograph of an array of elliptically polarized antennas at 94 GHz in a package. The generation of elliptically polarized waves Fig. 6 shows the set-up to generate a linearly polarized wave at any orientation. A linearly polarized wave at 375 GHz coming out of the open-ended waveguide through the polarizer was collimated by the lens. The wave passing through the half-wave plate was received by the modified bow-tie antennas at the back of the substrate lens. A half-wave plate is an anisotropic uniaxial Material [4]. The axes parallel and perpendicular to theopticalaxis have a slightly different dielectric constant. In our case it was made up of crystal quartz. The half-wave plate could rotate the electric field of the incident linearly polarized wave from 0 to 180 degrees. It was then picked up by antennas at the back of the substrate lens. Fig. 7 is a picture showing the set-up to generate a wave of any ellipticity or axial ratio. Linearly polarized waves at 94 GHz coming out of a horn were collimated by a lens. The wave was electronically 82 Polarized Antenna’s Response Fig. 3. Circularly polarized equiangular spiral antenna designed for use on a fused-quartz substrate (e, = 4). A 1,252; 0.04 X—+| -— eal $-317/2) ef ($-77) et $ e? (¢-7/2) Detector Antenna Low Frequency Lead a=0,175 83 Polarized Antenna’s Response in a DIP polarized antenna array Fig. 4. Circularly package. 84 Polarized Antenna’s Response Fig. 5. Elliptically polarized antenna array in a DIP package. 85 Polarized Antenna’s Response Fig. 6. Optical set-up to test the response of linearly polarized antennas to linearly polarized wave. ( A SUBSTRATE WITH ARRAY ANTENNA ON BACKSIDE OBJECTIVE LENS WIRE GRID POLARIZER ( , AQ wave, La { COLLIMATOR LENS CARCINOTRON OUTPUT FROM OPEN-ENDED WAVEGUIDE 86 Polarized Antenna’s Response modulated at 1 kHz. It then passed through a quarter- wave plate and an objective lens and received by antennas at the back of a substrate lens. Quarter-wave plate is like half-wave plate except that it is half its thickness. Depending on its orientation, it changes a linearly polarized wave from linear to elliptical to circular wave. The theoretical and experimental responses of the antennas to the waves The power received by any polarized antenna to any type of polarized wave was proportional to [5] a E 2 P. «= |E,*E,| (1) where E;, is the incident electric field vector and E, is the electric field excited by the antenna if it is transmitting. It is a unit vector. Both can be represented by Jones vectors [6]. The incident electric field is a linearly polarized wave going through either a half-wave or a quarter-wave plate. lLinearly polarized wave is represented as Cd 1 J. (2) The effect of the uniaxial crystal can be represented by [oor aaa | 0 \[ cosé | sine cosé@jlo e*%Jjl-sinée cose (3) where @ is the angle between the optical axis and the direction perpendicular to the linearly polarized wave 87 Polarized Antenna’s Response Fig. 7. Optical set-up to test the response of both the elliptically and circularly polarized antennas to elliptically polarized wave. . 88 Polarized Antenna'’s Response as shown in Fig. 8, and ¢ represents the amount of anisotropy of the crystal -~for quarter-wave plate, it is 90 degrees; for half-wave plate, it is 180 degrees. Now EY » the antenna effect, is 1 ] x (1 + §62)71/2 Set? (4) for linearly polarized antenna: & = 0 or infinite and y = 0 for circularly polarized antenna: 6 and y = tn/2. " = Putting equations 2,3,4 into 1 to get equation 5: Pr « sin?20(1-cos¢)/2+ 8 (sin26sin209(cosy-cos(¢-y))+ sin26cos76(cos(¢+y) (1-cos¢)-sin(¢+y7)sing))+ 82(14+(sin226/2) (-1+cosycos(y+¢)+sinysin(y+¢)). Equation 5 will be studied under three different conditions and verified by experiments. The three conditions all involved linearly polarized wave going through different uniaxial crystals and they are: (1). The response of a linear antenna to a linearly polarized wave with any orientation. A half-wave plate was used. (2). The response of a circular antenna to an elliptical wave. A quarter-wave plate was used. (3). The response of an elliptical antenna to an elliptical wave. 89 Polarized Antenna’s Response Fig. 8 The relationship between the incident electric field direction and the orientation of the uniaxial crystal. UNIAXIAL CRYSTAL INCIDENT ELECTRIC FIELD 90 Polarized Antenna’s Response Condition 1 : When the antenna is linear Phe 1-sin?26(1-cos¢)/2. If the uniaxial crystal has the exact thickness of a half-wave plate or ¢ = 180 degrees, then P, « cos720. Fig. 9 shows the experimental and theoretical results. The experiment was done using the set-up in Fig. 6 with the antenna shown in Fig. 2 and at 375 GHz. In the Millimeter wave area, the dielectric constants of materials sometimes are not well tabulated. So it would be difficult to know the exact thickness to make a half- wave plate or to make ¢ exactly 180 degrees. But from the theory, even if ¢ is 160 degrees, the response of the linearly polarized antenna will not be much affected. Condition 2: When the antenna is circular, with y = 1/2, P, « l-sin26@sing. If the uniaxial crystal has the exact thickness of a quarter-wave plate or ¢ = 90 degrees, then P, «= 1-sin2é6. Fig. 10 shows the theoretical and experimental results. The experimental work was done using the set-up in Fig. 7 and with the antenna shown in Fig. 4 and at 94 GHz. With ¢ = 110 degrees, the antenna response is still not 91 Polarized Antenna’s Response Fig. 9. The theoretical and experimental response of a linearly polarized antenna in the set-up shown in Fig. 6. q { | I —— THEORY ¢=180° aol ---- THEORY ¢=160° e EXPERIMENT 120 100 ANTENNA RESPONSE, nV re) fee) O oO B oO 20 6) I | 1 | o° 40° 80° 120° I60° RELATIVE POLARIZATION ANGLE 92 Polarized Antenna’s Response Fig. 10. The theoretical and experimental response of a circularly polarized antenna in the set-up shown in Fig. 7. 7 I -| | INCIDENT WAVE AXIAL RATIO or Li T tT Tf ae | | ee ee ee ee | 1 is . 4 oO aL L J Qu +- 4 oe ff FE LJ <¢ O.SP 4 Ow | re —— THEORY ¢ = 90° 1 — ---- THEORY ¢ = 110° = - e EXPERIMENT 4 O [ee ae SD SS MN z <x 93 Polarized Antenna’s Response much affected. Condition 3: When the antenna is elliptically polarized and the ‘uniaxial crystal has the exact thickness of a quarter- wave plate, then P. « gin?26(1-82)/2+ r 87-8sin26(sinyt+tcosycos20). Fig. 5 and Fig. 7 show the antenna and the set-up used for the experiment done at 94 GHz. The experimental points fit the theory if y = 75 degrees and & = 0.8 in equation 4. Fig. 11 shows the theoretical and experimental results. This chapter studies the response of any polarized antennas to any polarized waves. Experiments have been done in the millimeter wavelength. Results are close to theory. 94 Polarized Antenna's Response Fig. 11. The theoretical and experimental response of an elliptically polarized antenna in the set-up shown in Fig. 7. Ir ao = - LiJ = © -— _ ral QO a _ —— i & O.5b 4 Wo Ce = — gq =a az on - Li - S r — THEORY Lo e@ EXPERIMENT ° . e OLttt tt ttt td tt | -| [ INCIDENT WAVE AXIAL RATIO 4 95 Polarized Antenna’s Response Bibliography 1. P. P. Tong, D. P. Neikirk, P. E. Young, W. A. Peebles, N. C. Luhmann,Jr., D. B. Rutledge, "Imaging “polarimeter arrays for near-millimeter waves." IEEE Trans. Microwave Theory Tech., vol. MTT-32, pp. 507- 512, May 1984. 2. J. D. Dyson, "The Equiangular Spiral Antenna." IRE Trans. Antennas Propagat., vol. 7, pp. 181-187, 1959. 3. J. D. Kraus and K. R. Carver, Electromagnetics, 2nd edition . New York, McGraw-Hill, Chpt. 14. 4. F. A. Jenkins and H. E. White, Fundamentals of Optics, 4th edition. New York, McGraw-Hill, 1976, p. 567. 5. Y. C. Yeh, "The Received Power of a Receiving Antenna and the Criteria for its Design." Proc. of IRE, pp. 155-158, 1949. 6. E. Hecht and A. Zajac, Optics, 4th edition. Addison-Wesley, Chpt. 8. APP2/THE ANTENNA PATTERNS OF A LONG WIRE ANTENNA This appendix describes the antenna patterns of a long wire antenna on a dielectric at 94 GHz. Both theory and experiments will be shown. Fig. 1 is a photograph of a long wire antenna on a dielectric in a DIP package. Its width is 20 pm and its length is slightly more than 64,4 . Fig. 2 shows the tip of the antenna. The part of the line that is bright is Silver. The rough part is bismuth on silver while the bolometer detector is at the position as shown. The patterns of this antenna were measured at 94 GHz by the methods and set-up as described in Chapter 3. The theoretical and experimental H-plane patterns The theoretical H-plane or the plane with a transverse electric field or the ¢ direction in the spherical co- ordinate has already been done [1]. It was plotted out on Fig. 3 together with the experimental results for quartz substrate. The experimental results were close tothe theory. This, in fact, tested out the methods and set-up for this antenna pattern measurement technique. 97 Long Wire Antenna Fig. 1. A long wire antenna on a dielectric in a DIP package. 98 Long Wire Antenna Fig. 2. The tip of the long wire antenna. The bright line is silver. The rugged surface is bismuth on silver while the bismuth bolometer is at the position as shown. 99 Long Wire Antenna Fig. 3. The antenna patterns of a long wire antenna on dielectric. Only the patterns on the dielectric side are shown. The power radiated into the air side is at - least 10 dB down. | q oT } T T T T T ~ or + . H-PLANE 4 = a \ —— EXPERIMENT | 2 \. ---- THEORY Q ~~ uw ANTENNA 90° af Wt 05 Lo iy Cr <q — 2 = uJ - z < 0 0 0 T TIT T T T T ! or E- PLANE 7 s - ——_ EXPERIMENT ANTENNA—aJ 490° ~ Qa | ----paRAMETER- H a ow | FITTED j 7 | as 0 i << ost i tos Ww a ! aco <a 2 ; > LJ Ee 2 < 0 0 20 40 60 80 DEGREES 100 Long Wire Antenna The experimental and parameter-fitted E-plane pattern The E-plane pattern or the plane with a transverse _Magnetic field or the @ direction in the spherical co- ordinate of a long wire antenna on a dielectric will be shown below. The current along the wire is assumed to be of the forms: I = Cexp(-az-jpz). The far field E-plane radiation intensity of this current radiating into a material of dielectric constant n is proportional to [2]: K = | 1/((a/k)-j(nsino-p)) 1? with k as the free space wave number. The far field radiation intensity of this current ona dielectric material will be proportional to K multiplying the dipole radiation pattern on a dielectric (3]. For 0 < @ < the critical angle, it is proportional to: K x Mcos”6/(n*M+cos76+2ncoséM) with M = 1-n*sin?0. For the critical angle < @ < 90, it is proportional to: K x (-Mcos26)/Icos¢+jn(-M)1/2|2 0.23k 1.45k, With a and ww I 101 Long Wire Antenna for quartz substrate, the far field radiationintensity from the above equation is close to the experimental results. This is shown in Fig. 3. This gives us a feeling of both the attenuation and propagation constant of the wave in the wire antenna on a dielectric. The B -is the propagation constant of the wave along the wire. It is about 8% less than the propagation constant calculated from the quasi-static approximation [4]. This decrease is consistent with other measured values [5]. The a is the attenuation constant of the current along the wire. With the assumed value, the current will drop by 76% or 12.6 dB for every free space wavelength. This is a relatively large attenuation. 102 Long Wire Antenna Bibliography 1. N. Engheta, C. H. Papas and C. Elachi, “Interface Extinction and Subsurface Peaking of the Radiation - Pattern of a Line Source." Appl. Phys. B 26, pp. 231- 238, 1981. 2. S. Ramo, J. R. Whinnery and T. Van Duzer, Fields and Waves in Communication Electronics. John Wiley and Sons, 1965, Chpt. 12. 3. D. B. Rutledge and M. S. Muha, "Imaging Antenna Arrays." IEEE Trans Antennas Propagat., AP-30, pp. 535-540, 1982, 4. D. B. Rutledge, D. P. Neikirk and D. P. Kasilingam, "Integrated Circuit Antennas." Series on Infrared and Millimeter Waves, vol. 10, Academic Press, New York. 5. D. P. Kasilingam and D. B. Rutledge, "Surface-Wave Losses of Coplanar Transmission Lines.", IEEE/MfT-S Int. Microwave Symposium Digest, Boston, pp. 113-116, May 1983. ia eS GR eR CER AER GIR OURS RIG mi Sa Me shia entie APP3/THE COMPUTER PROGRAM FOR ANTENNA PATTERN MEASUREMENT 5 eis ci SU Si is RSE Si Ss SC ci SSA NSS li Ss oases 10 20 30 40 50 60 710 _ 80 90 100 110 120 130 140 150 160 170 180 190 200 210 220 230 240 250 260 270 280 290 300 310 320 330 340 350 360 370 380 390 400 410 420 “430 440 450 460 470 480 490 500° $10 320 REM REM REM REM REM REM REM REM REM REM REM REM REM REM REM REM REM REM REM REM REM 104 TAKE ANALOG DATA AT CHANNEL 3 AND ROTATE THE DATA ARE IN THE FILE CALLED “VDATA" THE FIRST SIX DATA ARE THETA: START ANGLE, RANGE AND STEP - AND THEN PHE : START ANGLE, RANGE AND STEP CLEAR ,14000 DEF SEG=7936 BLOAD "PCLAB",0 ANALOG ENTRY POINTS ADC.VALUE = 3 ADC.ON. TRIGGER=6 DIGITAL ENTRY POINTS ENABLE. FOR.OUTPUT=60 OUTPUT .DIGITAL.VALUE=66 OUTPUT.DIGITAL.ON. TRIGGER=72 SET. TIMEOUT=105 ANALOG INITIALIZATION HIGH.V! = +10! LOW.V! = -10! RANGE! = HIGH.V! - LOW.V! LSB!=RANGE! /65536! DIGITAL INITIALIZAION PORT. SELECT%=0 D.MASK%=s H1 TIMEOUTS=0 CALL ENABLE.FOR.OUTPUT (PORT.SELECT%) CALL SET.TIMEOUT (TIMEOUTS) D.STATE%=0 CALL OUTPUT.DIGITAL. VALUE THE A ARRAY FOR MOTOR ASCII CODES COMMANDS DIM A(10) "VDATA" IS THE RESULT DATA FILE. OPEN "VDATA" FOR OUTPUT AS #1 SET MOTORS VALUES XS=768 :REM X CONTROLLER STATUS ADDRESS YS=769 :REM Y CONTROLLER STATUS ADDRESS DR=770 :REM DATA REGISTER ADDRESS IX=771 :REM INTERRUPT RESET ADDRESS IY=772 :REM INTERRUPT RESET ADDRESS DS=773 :REM DUAL STATUS ADDRESS OUT OUT YS,190 :REM SET Y TO INITIAL VALUE XS,190 :REM SET X TO INITIAL VALUE SR=XS :REM SET DEFAULT CONTROLLER TO X y 1 | | 4 | | . i i | : L 530 $40 3350 560 370 380 590 600 610 620 630 640 650 660 670 680 690 700 710 720 730 740 150 760 170 780 790 800 810 820 830 840 850 860 870 880 885 890 900 910 920 930 -940 930 960 970 980 990 105 OUT SR,O :REM RESET CONTROLLER OUT SR,190 OUT IX,0: OUT IY,0 :REM CLEAR X,Y INTERRUPTS CLS INPUT “ENTER 1 FOR 1-D ROTATE AND 2 FOR 2-D"; FREEDOM IF FREEDOM=1 THEN GOTO 710 PRINT PRINT "FOR 2-D, ALWAYS FIX A THETA AND ROTATE PHE" CTR$="1" PRINT INPUT "ENTER THETA START ROTATION ANGLE"; TSTANG PRINT INPUT "ENTER THETA ROTATING RANGE IN DEGREE"; TRG PRINT INPUT “ENTER THETA ROTATING STEPS IN DEGREE"; TST PRINT INPUT "ENTER + FOR CW,- FOR CCW THETA ROTATION"; TDIR$ TST=TST*4 : GOTO 760 TSTANG=0 : TRG=0 : TST=4 : TT$="0" : TDIR$="—" PRINT INPUT “ENTER 1 FOR PHE OR 2 FOR THETA ROTATION"; CTR$ PRINT PRINT "CALL THE ROTATION AS PHE" CLS : INPUT "ENTER PHE START ROTATION ANGLE"; STANG PRINT : INPUT "ENTER PHE ROTATION RANGE IN DEGREE ";RG PRINT : INPUT "ENTER PHE ROTATION STEPS IN DEGREE"; ST ST=ST*4 WRITE #1,TSTANG,TRG,TST,STANG,RG,ST REM generate character equiv. of ST and put it in S$ CHAR=ST GOSUB 1850 S$=CHA$ REM GENERATE CHARACTER EQUIV. OF TST,RESULT IN TT$ CHAR=TST GOSUB 1850 TT$=CHA$ PRINT INPUT "ENTER + FOR CW,.- FOR CCW PHE ROTATION" ; DIR$ B$=CTR$ GOSUB 1720 BS="R 180" GOSUB 1480 B$S="§ 35" GOSUB 1480 B$="F 9" GOSUB 1480 B$="A" : GOSUB 1480 FOR TVAR=1 TO 4*TRG/TST+1 STEP 1 1000 B$=CTR$ 1010 GOSUB 1720 1020 REM 1030 B$=DIR$ : GOSUB 1480 106 1040 REM LOOP TO TAKE A DATA, WAIT AND THEN ROTATE 1050 FOR VAR=1 TO 4*RG/ST STEP 1 1060 GOSUB 1980 1070 WRITE #1,V - 10980 SS$=S$ 1090 SST=ST 1100 § GOSUB 1330 1110 NEXT VAR : GOSUB 1980 : WRITE #1,V 1120 REM rotate motor back to the starting position 1130 REM generate the character equivalent of rg*4 1140 CHAR=RG*4 1150 GOSUB 1850 1160 SS$=CHA$ 1170 IF DIR$="-" THEN VDIR$="+" ELSE VDIR$="—-" 1180 B$=VDIR$ 1190 GOSUB 1480 1200 SST=RG*4 1210 GOSUB 1330 1220 B$="2" ; GOSUB 1720 1230 B$=TDIR$ : GOSUB 1480 1240 SS$=TT$ 1250 SST=TST 1260 GOSUB 1330 | a 4 a. . : | iy 2 1 4 1350 NEXT K 1270 NEXT 1280 REM de-energize the stepper motor 1290 B$="I"" 1300 GOSUB 1480 1310 REM PROGRAM ENDS 1320 GOTO 2390 1330 REM stepper motor rotation control 1340 B$="E" 1350 GOSUB 1480 1360 B$="N "+SS$ 1370 GOSUDB 1480 1380 BS="G" 1390 GOSUB 1480 1400 B$="0" 1410 GOSUB 1480 1420 BS="Q" 1430 GOSUB 1480 : B$="D" : GOSUB 1480 1440 IF SST<20 THEN WSTEP=6*20 ELSE WSTEP=6*ST 1450 REM PROGRAM WAIT TILL MOTOR IS STEADY | ‘1460 FOR DEL=1 TO WSTEP : WASTE =LOG(DEL) : NEXT DEL [ 1470 RETURN i 1480 C=0 | 1490 J=LEN(B$) i 1500 FOR K=1 TO J | 1510 C=C+1 . 1520 S=ASC(MID$(B$,K,1)) / 1530 IF S=59 THEN S=44 | 1540 A(C)=S 107 1560 IF A(1)=81 GOTO 1590 1570 C=C+1 1580 A(C)=13 1590 OUT SR,190 1600 FOR K=1 TO C 1610 S=INP(SR) 1620. IF §¢128 GOTO 1610 1630 OUT DR,A(K) 1640 S=INP(DR) 1650 IF S)<A(K) GOTO 1810 1660 OUT SR,62 1670 S=INP(SR) 1680 IF S)=128 GOTO 1670 1690 OUT SR,190 1700 NEXT K 1710 RETURN 1720 REM subroutine of the motor control / 1730 OUT XS,0: OUT YS,0: OUT DS,0 | 1740 ON VAL(B$) GOTO 1750,1760,1770 | 1750 OUT XS,190: SR=XS: GOTO 1800 | 1760 OUT YS,190: SR=¥S: GOTO 1800 / 1770 OUT DS,190: SR=DS | 1780 IF SR=XS GOTO 1800 | 1790 IF SR=YS GOTO 1800 | 1800 RETURN 1810 PRINT "data error: program aborted" | 1820 GOTO 2400 1830 REM | 1840 REM 1850 REM GENERATE THE CHARACTER EQUIVALENT OF CHAR 1860 REM PUT IT IN CHA$ 1870 REM 1st SIGNIFICANT DIGIT ' 1880 S1=CHAR/100 : S1=FIX(S1) i 1890 REM 2nd SIGNIFICANT DIGIT | 1900 S2=(CHAR-S1*100)/10 : S2=FIX (S2) i 1910 REM 3rd SIGNIFICANT DIGIT | 1920 S3=FIX( CHAR-S1*100-52*10) | 1930 CHA$="" i 1940 IF CHAR > 99 THEN CHA$=CHR$(S1+48) i 1950 IF CHAR >9 THEN CHA$=CHA$+CHR$(S2+48) | 1960 CHA$ =CHA$+CHR$(S3+48) | 1970 RETURN | 1980 REM i 1990 REM ANALOG DATA ACQUISITION l 2000 REM i 2010 G=4 i 2020 CHANNELS%=3 [ 2030 GOSUB 2100 | 2040 REM [ 2050 REM DIGITAL OUTPUT | 2060 REM 2070 D.STATE%=1 avaisessaarsine 108 2080 CALL OUTPUT.DIGITAL. VALUE 2090 RETURN 2100 REM 2110 REM SET VOLTAGE RANGE 2120 REM 2130 ON G GOTO 2140,2170,2220,2270 2140. GAIN®%=500 2150 GOSUB 2310 2160 IF V<.0199 THEN G=4: RETURN 2170 GAIN%=100 2180 GOSUB 2310 2190 IF (.0199<V) AND (V<.0999) THEN G=3: RETURN 2200 IF V>.0999 THEN GOTO 2220 2210 GOTO 2140 2220 GAIN%=10 2230 GOSUB 2310 2240 IF (,0999<V) AND (V<.99) THEN G=2: RETURN 2250 IF V>.99 THEN GOTO 2270 2260 GOTO 2170 2270 GAIN®S=1 2280 GOSUB 2310 2290 IF V>.99 THEN G=1: RETURN 2300 GOTO 2220 2310 REM 2320 REM MEASURE VOLTAGE 2330 REM 2340 SCALED. LSB!=LSB!/GAIN%S 2350 CALL ADC.VALUE (CHANNEL®, GAIN®, ANALOG. VALUE®) 2360 VOLTAGE! = (ANALOG.VALUE® * SCALED.LSB!) 2370 V=ABS (VOLTAGE! ) 2380 RETURN 2390 CLOSE #1 2400 CLS 2410 END